Boltzmann machine, method for controlling boltzmann machine, and information processing device having boltzmann machine

ABSTRACT

Boltzmann machine includes a plurality of circuit units each having an adder that adds weighted input signals and a comparison unit that compares an output signal of the adder with a threshold signal to output a binary output signal; and digital arithmetic units each generating the weighted input signals by weighting the binary output signal of the circuit units with a weight. The comparison unit has a first comparator that compares a thermal noise with a reference voltage to output a binary digital random signal, a DA converter that converts the digital random signal to an analog random signal and varies a magnitude of the analog random signal, and a second comparator that compares the output signal of the adder with the analog random signal to generate the binary output signal with a predetermined probability function.

CROSS-REFERENCE TO RELATED APPLICATION

This application is based upon and claims the benefit of priority of the prior Japanese Patent Application No. 2016-018668, filed on Feb. 3, 2016, the entire contents of which are incorporated herein by reference.

FIELD

The present invention relates to a Boltzmann machine, a method for controlling the Boltzmann machine, and an information processing device having the Boltzmann machine.

BACKGROUND

A Boltzmann machine as a type of neural network is expected to be capable of processing multivariable optimization and pattern recognition at high speed. The concept of the Boltzmann machine is that the configuration of a network is held equivalent to a thermodynamic system and it is ensured that an energy state reaches a thermal equilibrium state when sufficient time has elapsed, according to a Boltzmann distribution.

In relation to the Boltzmann machine, the Ising model is known that is a functional unit for determining an optimal solution stochastically and has a plurality of lattice points each having a spin status and links between the lattice points. Examples thereof are disclosed in Japanese Laid-open Patent Publication No. H2-27493 and Japanese Laid-open Patent Publication No. H4-18661. The Ising model is a lattice model that is constituted by the lattice points each having two orientation statuses and considers an interaction only with the adjacent lattice point. Japanese Laid-open Patent Publication No. 2014-96769 and Japanese Laid-open Patent Publication No. 2013-143626 are related prior arts.

SUMMARY

In the Boltzmann machine, in order to prevent the solution from falling into a local minimum value, a method called simulated annealing is used which, using a temperature parameter, changes a probability with which each lattice (unit) outputs+1.

In the conventional Boltzmann machine, in order to change the above probability, a noise is generated by a random noise generator, the amplitude of the generated noise is changed by an amplifier, and the output value of each lattice (unit) is thereby generated. With this, the probability that each lattice (unit) outputs+1 is changed using the temperature parameter. However, the circuit scale of the random noise generator is large and, since the magnitude of the generated noise is random, it is difficult to appropriately change the amplitude of the noise.

According to an aspect of the embodiments, Boltzmann machine includes a plurality of circuit units each of which has an adder that performs addition of a plurality of weighted input signals and a comparison unit that compares an output signal of the adder with a threshold signal to output a binary output signal; and a plurality of digital arithmetic units each of which generates each of the weighted input signals by weighting the binary output signal of a first circuit unit of a pair of the circuit units in the plurality of circuit units with a weight to output the weighted input signal to a second circuit unit of the pair of the circuit units, wherein the comparison unit has: a first comparator that compares a thermal noise with a reference voltage to output a binary digital random signal; a first DA converter that converts the digital random signal to an analog random signal, and varies a magnitude of the analog random signal; and a second comparator that input the analog random signal as the threshold signal, and compares the output signal of the adder with the threshold signal to generate the binary output signal with a predetermined probability function.

According to the above aspect, it is possible to vary the amplitude of the random noise with a small-scale circuit.

The object and advantages of the invention will be realized and attained by means of the elements and combinations particularly pointed out in the claims.

It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are not restrictive of the invention.

BRIEF DESCRIPTION OF DRAWINGS

FIGS. 1A and 1B are views illustrating an example of the Ising model.

FIGS. 2A and 2B are views illustrating the Ising model of an undirected graph of three lattice points and a network circuit corresponding thereto.

FIGS. 3A-3C are views illustrating the annealing of the Boltzmann machine.

FIG. 4 is a view illustrating the sigmoid function.

FIG. 5 is a view illustrating the arithmetic circuit of the Ising model in the present embodiment.

FIGS. 6A and 6B are views illustrating an example of the circuit of the comparator and the operation thereof.

FIG. 7 is a view illustrating the configuration of the comparator (comparison unit) dn.

FIG. 8 is a view illustrating the signal waveform of the operation of the first comparator.

FIG. 9 is a view illustrating a sampling operation of the first DA converter dndac.

FIG. 10 is a view illustrating a hold operation of the first DA converter dndac.

FIG. 11 is a view illustrating a logical value table of the operation of a control circuit 10 in the first DA converter dndac.

FIG. 12 is a view illustrating the signal waveform of the operation of each of the first DA converter dndac and the second comparator dn2.

FIGS. 13A-13C are views illustrating the output function of the second comparator.

FIG. 14 is a view illustrating the configuration of the comparison circuit (comparison unit) in a second embodiment.

FIG. 15 is a view illustrating the signal waveform of the comparison circuit (comparison unit) in the second embodiment.

FIGS. 16A-16C are views illustrating the output function of the second comparator in the case where the number of pairs of the digital random signal generators and the first DA converters is one, two, or three.

FIG. 17 is a flowchart illustrating the operation of the Boltzmann machine in the present embodiment.

FIG. 18 is a flowchart illustrating the detail of the execution step S4 of the Boltzmann machine.

FIGS. 19A-19C are views illustrating three phases of the execution of the Boltzmann machine in the present embodiment.

FIG. 20 is a view illustrating an information processing device that has the Boltzmann machine.

DESCRIPTION OF EMBODIMENTS

FIGS. 1A and 1B are views illustrating an example of the Ising model. FIG. 1A illustrates an example of the configuration of a model that represents the behaviors of magnetic spins, and FIG. 1B illustrates a graph that represents the relationship between energy E of a network and a combination pattern of the spin “Spin status”.

The Ising model in FIG. 1A is a model that represents the behaviors of the magnetic spins, and is constituted by nine lattice points each having two orientation statuses. When it is assumed that each of the two orientation statuses of each lattice point is a spin q, an interaction coefficient between the adjacent lattice points, i.e., a weight of coupling is w, and an external magnetic field coefficient, i.e., the force of a magnetic field given to each lattice point from the outside is b, the formula of the energy of the network is defined by the following Math. 1.

$\begin{matrix} {E = {{- {\sum\limits_{{< i},{j >}}{w_{ij}q_{i}q_{j}}}} - {\sum\limits_{j}{b_{j}q_{j}}}}} & \left\lbrack {{Math}.\mspace{14mu} 1} \right\rbrack \end{matrix}$

Herein, the value of the spin q is +1 when the orientation of the spin is an upward orientation and is −1 when the orientation thereof is a downward orientation, and a spin q_(i) of a given lattice point is subjected to an interaction of a spin q_(j) of the adjacent lattice point with the weight of an interaction coefficient w_(ij). In addition, in Math. 1, <i, j> means a combination of i and j and, e.g., <1, 2> and <2, 1> are not duplicated because there are the same combination.

In the graph in FIG. 1B, the horizontal axis indicates the combination pattern of the spin, the vertical axis indicates energy E of the network, and a point that minimizes the energy (Solution) indicates an optimal state. By determining the pattern of the spin at the point that minimizes the energy, it is possible to obtain an optimal solution.

Next, a description will be given of a method for determining the optimal solution at the point that minimizes the energy in the case where the formula in Math. 1 above is defined as the energy of the network.

FIGS. 2A and 2B are views illustrating the Ising model of an undirected graph of three lattice points and a network circuit corresponding thereto. For simplifying the description, consideration will be given to a model that has three lattice points (nodes) and interactions (links) illustrated in FIG. 2A.

A node i has a spin status q_(i) and a bias b_(i) (q_(i), b_(i)), and i=1, 2, or 3 is satisfied. The link has a weight w and, since the model is an undirected model, the weight of the link that connects the node i and a node j is w_(ij), and a link weight w_(ii) to the node i satisfies w_(ii)=0. In addition, the weight w_(ij) that acts on the link _(j) by the link i is equal to a weight w_(ji) that acts on the link i by the link j.

Herein, according to Math. 1 described above, the energy of the network is defined. The energy E in Math. 1 in the model of three lattice points in FIG. 2A is as follows: E=−(w ₁₂ q ₁ q ₂ +w ₁₃ q ₁ q ₃ +w ₂₃ q ₂ q ₃)−(b ₁ q ₁ +b ₂ q ₂ +b ₃ q ₃).  [Math. 2]

New energy E′ of the network in the case where only the spin q₁ of the node 1 is inverted and changed to a new spin q′₁ (=−q₁) is as follows: E′=−(w ₁₂ q′ ₁ q ₂ +w ₁₃ q′ ₁ q ₃ +w ₂₃ q ₂ q ₃)−(b ₁ q′ ₁ +b ₂ q ₂ +b ₃ q ₃).  [Math. 3]

Consequently, a change amount ΔE=E′−E of the energy when only the spin q₁ of the node 1 is inverted and changed to the new spin q′₁ (=−q₁) is as follows, by simplifying the calculation formula of ΔE based on q′₁=−q₁:

$\begin{matrix} \begin{matrix} {{\Delta\; E} = {{E^{\prime} - E} = {{{- w_{12}}q_{1}^{\prime}q_{2}} - {w_{13}q_{1}^{\prime}q_{3}} + {w_{12}q_{1}q_{2}} +}}} \\ {{w_{13}q_{1}q_{3}} - {b_{1}q_{1}^{\prime}} + {b_{1}q_{1}}} \\ {= {{- 2}{q_{1}^{\prime}\left( {{w_{12}q_{2}} + {w_{13}q_{3}} + b_{1}} \right)}}} \end{matrix} & \left\lbrack {{Math}.\mspace{14mu} 4} \right\rbrack \end{matrix}$

When the formula in Math. 4 described above is generalized based on q₁=q_(i), the formula is as follows: ΔE=−2q′ _(i)(Σ_(j) w _(ij) q _(j) +b _(i)).  [Math. 5]

j is all numbers except for i.

From the formula of the change amount ΔE of the energy in Math. 5, it can be said that, when the sign of the new spin status q′_(i) of a given node i matches the sign of Σ_(j)w_(ij)q_(j)+b_(i) determined in the original spin status, the energy of the network is reduced. From this, by configuring an arithmetic circuit that causes the sign of the new spin status q′_(i) of the given node i to match the sign of Σ_(j)w_(ij)q_(j)+b_(i) determined in the original spin status, it is possible to gradually reduce the energy E of the network and eventually obtain the smallest value that satisfies ΔE=0. The combination pattern of the spins q_(i) of the individual nodes when the smallest value of the energy E is obtained is the optimal solution of the model to be determined.

FIGS. 2A and 2B illustrate an arithmetic circuit that reduces the network energy E described above for the model of three nodes NODE 1-3. In the arithmetic circuit, the three nodes are constituted by adders a1, a2, and a3 that weight the spins q of the adjacent nodes with the interaction coefficient w between the nodes and perform addition of the spins q, and comparators d1, d2, and d3 that compare outputs z1, z2, and z3 of the adders with values of biases b1, b2, and b3 (to be accurate, values obtained by inverting the sign of the bias), and output the spins q. The individual comparators d1, d2, and d3 operate with different clocks.

For example, in the adder d1 of the node 1, the following addition is performed: z ₁ =w ₁₂ q ₂ +w ₁₃ q ₃  [Math. 6]

Therefore, the adder of each node performs the following addition in general: z _(i)=Σ_(j) w _(ij) q _(j)  [Math. 7]

A comparator d_(i) of the arithmetic circuit compares an addition result z_(i) with the value of −b_(i) obtained by inverting the sign of a bias b_(i), and outputs a comparison result q′_(i)=+1 or −1. That is, the operation of the comparator is as follows:

the comparator d_(i) outputs q′_(i)=+1 when z_(i)>−b_(i), and the comparator d_(i) outputs q′_(i)=−1 when z_(i)<−b_(i).

When the above inequalities are modified, the operation of the comparator is as follows:

the comparator d_(i) outputs q′_(i)=+1 when z_(i)+b_(i)>0, and

the comparator d_(i) outputs q′_(i)=−1 when z_(i)+b_(i)<0.

According to Math. 7, the left side of the inequalities is as follows: z _(i) +b _(i)=Σ_(j) w _(ij) q _(j) +b _(i)

Therefore, it follows that the comparator d_(i) of the arithmetic circuit performs the following comparison operation:

the comparator d_(i) outputs q′_(i)=+1 when Σ_(j)w_(ij)q_(j)+b_(i)>0, and

the comparator di outputs q′_(i)=−1 when Σ_(j)w_(ij)q_(j)+b_(i)<0.

That is, this means that the comparator d_(i) implements the match between the sign of q′_(i) and the sign of Σ_(j)w_(ij)q_(j)+b_(i) as the condition for the reduction of the energy change ΔE, and hence the state of the smallest value of the energy E of the network (a state in which ΔE=0 is satisfied) can be determined by the arithmetic circuit in FIG. 2. That is, the operation of the comparator d_(i) and the change of the energy E are as follows:

when Σ_(j)w_(ij)q_(j)+b_(i)>0, since the comparator d_(i) outputs q′_(i)=+1, both of the signs are positive signs and the energy is reduced,

when Σ_(j)w_(ij)q_(j)+b_(i)<0, since the comparator d_(i) outputs q′_(i)=−1, both of the signs are negative signs and the energy is reduced, and,

when the operation is repeated until ΔE=Σ_(i) {−2q′_(i)(Σ_(j)w_(ij)q_(j)+b_(i))}=0 is satisfied, the energy E reaches the smallest value. Alternatively, by repeating the operation, ΔE approaches 0, and the energy E reaches a value close to the smallest value.

Consequently, in the arithmetic circuit in FIG. 2, when the operation for inverting the status of the spin q of a given node is repeated randomly, the change amount of the energy approaches ΔE=0 in a certain time, and the energy E can reach the state having the value close to the smallest value.

However, when the spin q′_(i) is definitely determined, it is not possible to escape from a local solution. To cope with this, by introducing the operation for stochastically determining the spin q′_(i) into the comparator d_(i), the comparator d_(i) is allowed to operate so as to increase the energy, and it becomes possible to escape from the local solution and reach the minimum value (optimal solution) close to the smallest value. By using an annealing method in which the stochastic degree shifts from a high state to a low state, the energy can be caused to reach the minimum value close to the smallest value.

Specifically, as will be described later, by adding a noise to a threshold −b_(i), the comparator d_(i) is caused to perform an operation for outputting q′_(i)=−1 even when Σ_(j)w_(ij)q_(j)+b_(i)>0, and outputting q′_(i)=+1 even when Σ_(j)w_(ij)q_(j)+b_(i)<0, and the energy E is thereby increased.

That is, when the comparators d_(i) of the three nodes are caused to operate in random order, every time an operation for causing the sign of Σ_(j)w_(ij)q_(j)+b_(i) to match the sign of q′_(i) is performed, the energy E is reduced. However, by adding the noise to the threshold −b_(i), the comparator d_(i) malfunctions, and increases the energy E. By the operation that increases the energy, the energy is allowed to escape from the local solution.

FIG. 3 is a view illustrating the annealing of the Boltzmann machine. As a method for deriving the optimal solution that minimizes the energy E, there is a method that determines the solution by the annealing (a method called annealing in metallurgical engineering). In the annealing, by slowly cooling metal or glass after heating the metal or glass to an appropriate temperature, an internal distortion thereof is removed. When the temperature of heat is high, atoms remain in a high-energy state but, by the slow cooling, the atoms transition such that the energy becomes lower than that in an initial state, and crystals are stabilized.

The horizontal axis in FIG. 3 corresponds to the combination pattern of the spin q, and the vertical axis therein corresponds to the energy E. In addition, a curve represents the energy E of the network, and a point MIN represents the smallest energy and is the optimal solution. The minimum value of each of points other than the point MIN is the local minimum value of the energy, and indicates the local solution. Normally, in a method such as a steepest descent method or the like, the energy does not move so as to be higher, and hence, once the energy falls into the local solution, it is not possible to escape from the local solution. However, in the Boltzmann machine, by addition of a thermal noise, the energy can be stochastically changed so as to be higher to a certain extent, therefor, the energy can escape from the local solution.

In FIG. 3, the range of the changeable energy is indicated by e_(var). The range of the changeable energy e_(var) is illustrated conceptually, and is not directly related to the curve indicative of the energy E. In FIG. 3, when the temperature is high as in FIG. 3A, since the thermal noise is large, the range of the changeable energy e_(var) is maximized, the optimal solution MIN is included, and it is possible to escape not only from the local solution but also even from the optimal solution MIN. When the temperature is reduced and the thermal noise is reduced to be medium, as illustrated in FIG. 3B, the range of the changeable energy e_(var) become a medium range, the optimal solution MIN is not included, and it is possible to escape from other local solutions although it is not possible to escape from the optimal solution MIN. When this state is continued, a possibility that the optimal solution MIN is achieved is increased rapidly. Further, when the temperature is reduced and the thermal noise is minimized, as illustrated in FIG. 3C, the range of the changeable energy e_(var) is minimized, and the transition of the energy gradually ceases.

Consequently, in the arithmetic circuit in FIG. 2, when the degree of the stochastic operation of each of the comparators d1, d2, and d3 is gradually reduced while the operation for inverting the status of the spin q of a given node is repeated randomly, the energy E of the network can be expected to reach the optimal solution MIN as the smallest value (or a suboptimal solution of a value close to the smallest value) after the repetition of the predetermined number of times. This control of the degree of the stochastic determination is implemented by controlling the gradient of a sigmoid function as an output function of the comparators d_(i) using the magnitude of the thermal noise.

FIG. 4 is a view illustrating the sigmoid function. The horizontal axis corresponds to an input x (an input z of the comparator d_(i)), and the vertical axis corresponds to a probability f(x) that the output q of the comparator d_(i) is +1. FIG. 4 illustrates that the probability f(x) that the output q is +1 when the input x of the comparator is compared with a threshold 0 (herein, the threshold b=0) approaches 1.0 in the case where the input x is larger than the threshold 0, and approaches 0.0 in the case where the input x is smaller than the threshold 0. This function f(x) is the sigmoid function. The threshold 0 corresponds to the threshold b of the comparator d_(i).

The sigmoid function is often used for representing the characteristic of the annealing. As described above, the comparator d_(i) receives the input x and, when the input x exceeds the threshold b (b=0 in FIG. 4), the comparator d_(i) outputs +1 with a given possibility f(x). The function f(x) is illustrated in FIG. 4 and, x represents an input value, and T represents a temperature parameter.

FIG. 4 illustrates three graphs with the temperatures T of 0.5, 1, and 2. When the temperature is high like T=2, the gradient of the function f(x) is gradual and, even in the case where the input x deviates from the threshold 0, the probability that the output is +1 is gradually increased or reduced. This state corresponds to FIG. 3A described above. When the temperature T is gradually reduced, the gradient of the function f(x) gradually becomes steep and, in the case where the input x deviates from the threshold 0, the probability that the output is +1 immediately becomes 1.0 or 0.0. This corresponds to FIG. 3B or FIG. 3C described above. Thus, by implementing the change of the characteristic of the sigmoid function f(x) on the circuit, the annealing is allowed.

Present Embodiment

Hereinbelow, the arithmetic circuit of the Ising model of the present embodiment will be described. FIG. 5 is a view illustrating the arithmetic circuit of the Ising model in the present embodiment. The arithmetic circuit in FIG. 5 has a configuration obtained by generalizing the arithmetic circuit for the model of three nodes illustrated in FIG. 2. That is, the arithmetic circuit in FIG. 5 corresponds to the Ising model having the node 1 to the node n illustrated in FIG. 1. Note that FIG. 5 illustrates the node 1 and the node n only, and the configuration of the remaining nodes is omitted.

A circuit unit of a node ND_1 has a digital adder a1 that performs addition of q_(i)*w_(i1) obtained by multiplying the spin q_(i) of the adjacent node that exerts the interaction on the node ND_1 by the corresponding weight w_(i1), a DA converter c1 that performs DA conversion of a digital output z_(1d) of the digital adder, and a comparator d1 that compares an analog output z₁ with the threshold b_(i) and outputs the spin q₁. When the threshold b1 is assumed to be 0, the spin q₁ is +1 or −1. Similarly to the node ND_1, the circuit unit of a node ND_n has a digital adder an, a DA converter cn, and a comparator (comparison unit) dn.

In addition, the arithmetic circuit has a digital arithmetic unit e_(ij) that multiplies the spin q_(i) as the output of each node by the corresponding weight w_(ij) (e_(1n) and e_(n1) are illustrated in FIG. 5).

FIG. 5 illustrates the detailed configuration of the comparator (or the comparison unit) dn of the node ND_n. The comparator (or the comparison unit) dn has a first comparator dn1, a first DA converter dndac, and a second comparator dn2. The first comparator dn1 compares a thermal noise Voi with a reference voltage Vnb, and outputs a binary digital random signal (Vdran). The first DA converter dndac converts the digital random signal Vdran to an analog random signal Ve, and varies the magnitude of the analog random signal Ve based on an amplitude control signal Nvar. The second comparator dn2 adds the analog random signal Ve to a threshold signal bn and input the resulting signal bn+Ve, compares a threshold signal Vo (=bn+Ve) with an output signal z_(n) of the digital adder an, and generates a binary output signal q_(n) with a predetermined probability function.

A constant voltage Vnb (=GND) is input to the positive input of the first comparator dn1, and a noise voltage Vnoi of the thermal noise generated in a resistor Rnoi is input to the negative input of the first comparator dn1. A capacitance Cnoi is, e.g., a parasitic capacitance of the negative input terminal. In addition, the amplitude control signal Nvar is an m-bit digital amplitude control signal. An analog output z_(n) of the second DA converter cn, which is obtained by converting a digital output z_(nd) of the digital adder an, is input to the positive input of the second comparator dn2, and the threshold signal Vo=bn+Ve is input to the negative input of the second comparator dn2.

FIGS. 6A, 6B are views illustrating an example of the circuit of the comparator and the operation thereof. FIG. 6A illustrates the example of the circuit of the first comparator dn1 or the second comparator dn2 on the left side. The comparator is a dynamic latch comparator that operates in synchronization with a clock CK. The clock CK is supplied to a gate of a switch SW14, an inverted clock XCK is supplied to each of control terminals of switches SW10, SW11, SW12, and SW13, and each of the switches is turned ON when the clock or the inverted clock is at an H level, and is turned OFF when the clock or the inverted clock is at an L level.

The comparator has an NMOS transistor N5 to the gate of which a positive input signal IP is supplied, and an NMOS transistor N6 to the gate of which a negative input signal IM is input, and has a switch SW14 between a ground GND and a source SS that are shared by the transistors N5 and N6. Further, the comparator has PMOS and NMOS transistor pairs P1 and N3 and P2 and N4 between drains DN and DP of the transistors N5 and N6 and a power source voltage VDD, and gates and drains of the transistor pairs P1 and N3 and P2 and N4 are cross-connected to configure a latch circuit. In addition, drains of the transistor pairs P1 and N3 and P2 and N4 are connected to a negative output ON and a positive output OP respectively.

When the clocks CK=L and XCK=H are satisfied, the switches SW10, SW11, SW12, and SW13 are turned ON and the switch SW14 is turned OFF, and the comparator is reset. In the reset state, the drain nodes DN and DP are connected to the power source voltage VDD, and the output terminals ON and OP are also connected to the power source voltage VDD. When the clocks CK=H and XCK=L are satisfied, the switches SW10, SW11, SW12, and SW13 are turned OFF and the switch SW14 is turned ON, and the input transistors N5 and N6 compare the input signals OP and IM with each other. In the case where IP>IM is satisfied, the transistor N5 is brought into conduction more strongly than the transistor N6, the reduction of the drain node DN is larger than that of the drain node DP, and DN<DP is satisfied. Subsequently, this state is latched by the latch circuit constituted by the transistor pairs P1 and N3 and P2 and N4, and the output signals satisfy ON<OP. In the case where IP<IM is satisfied, the above operation is reversed.

The comparison circuit in FIG. 6 is only exemplary, and the comparison circuit having another configuration may also be used.

FIG. 7 is a view illustrating the configuration of the comparator (comparison unit) dn. The configuration is the same as the configuration of the comparator in FIG. 5 but, in an example in FIG. 7, a clock CK1 is supplied to the first comparator dn1, a clock CK2 is supplied to the second comparator dn2, and, in addition to the amplitude control signal Dvar, a sample clock CKsam and positive and negative reference voltages +Vref and −Vref are input to the DA converter dndac. The DA converter dndac is a capacitance DAC that uses capacitative elements, and the capacitance value of the capacitive elements is varied and the amplitude of the analog output signal Ve is varied with the amplitude control signal Dvar.

Next, a description will be given of the operation for generating the digital random signal Vdran based on the thermal noise Vnoi by the first comparator dn1.

FIG. 8 is a view illustrating the signal waveform of the operation of the first comparator. When the thermal noise is generated in the semiconductor region of the resistor Rnoi connected to the negative input of the first comparator dn1, charges flow into the parasitic capacitance Cnoi, and the thermal noise voltage Vnoi fluctuates. That is, the resistor Rnoi and the capacitance Cnoi constitute a thermal noise generation circuit. The thermal noise is generated at random timings, and hence the thermal noise voltage Vnoi fluctuates randomly.

The first clock CK1 is input to the first comparator, and the first comparator dn1 outputs the comparison result of the thermal noise voltage Vnoi and the constant voltage Vnb as the digital random signal Vdran that has +1 (Vnb>Vnoi) or −1 (Vnb<Vnoi) at the rising edge of the first clock CK1. Thus, the digital random signal Vdran is the sign of +1 or −1, and is a sign noise that changes randomly. As described above, the first comparator dn1 converts the weak analog random noise Vnoi to the digital random signal Vdran.

FIGS. 9 and 10 are views illustrating a sampling operation and a hold operation of the first DA converter dndac. FIG. 11 is a view illustrating a logical value table of the operation of a control circuit 10 in the first DA converter dndac. In addition, FIG. 12 is a view illustrating the signal waveform of the operation of each of the first DA converter dndac and the second comparator dn2.

The first DA converter dndac illustrated in each of FIGS. 9 and 10 has m capacitances 1C (to be accurate, a capacitor having capacitance 1C, hereinafter referred to as the capacitance 1C) each having one electrode connected to the node at which the analog random signal Ve is generated, and switches SW_1 to SW_m connected to the other electrodes of the capacitances 1C respectively. Each of the switches SW_1 to SW_m connects the node of the reference voltage +Vref, the reference voltage −Vref, or a ground voltage 0 volt to the other electrode of each of the capacitances 1C based on an m-bit control signal p[m]. The control circuit 10 receives the m-bit amplitude control signal Dvar[m] that controls the amplitude of the noise, the digital random signal Vdran, and the sample clock CKsam, and generates the control signal p[m] for each of the switches SW_1 to SW_m. In the drawing, a capacitance Cc is an input terminal capacitance of the second comparator dn2, and the capacitance value thereof is very small as compared with that of the capacitance 1C.

As illustrated in FIG. 9, when the sample clock CK_(sam) is in a sampling mode in which CK_(sam)=1 is satisfied, a control signal p_(sam) of a switch SW1 satisfies p_(sam)=1, the switch SW1 is turned ON, and the threshold voltage bn is sampled in all of the capacitances 1C. The control circuit 10 controls the m-bit control signals p[m] such that the switches SW_1 to SW_m are connected to the ground 0 in synchronization with CK_(sam)=1. With this, the threshold voltage bn is applied to all of the capacitances C1, and the voltages thereof are sampled. During a sampling period, an input voltage Vo of the second comparator satisfies Vo=bn.

Next, as illustrated in FIG. 10, when the sample clock CK_(sam) is in a hold mode in which CK_(sam)=0 is satisfied, the control signal p_(sam) of the switch SW1 satisfies p_(sam)=0, the switch SW1 is turned OFF, the node to which the analog random signal Ve is output or the node of the input voltage Vo of the second comparator dn2 is brought into a floating state.

As indicated by the logical value table in FIG. 11, the control circuit 10 controls the control signal p[m] as +Vref or −Vref based on the fact that, the digital random signal Vdran is +1 or −1, when the m-bit-th amplitude control signal Nvar[m] is 1. That is, the switch SW_m is connected to the positive reference voltage +Vref or the negative reference voltage −Vref when the control signal p[m] is +Vref or −Vref. That is, when the amplitude control signal Nvar[m] is 1, the switch SW_m is connected to the positive reference voltage +Vref in the case where the digital random signal satisfies Vdran=+1, and the switch SW_m is connected to the negative reference voltage −Vref in the case where the digital random signal satisfies Vdran=−1.

On the other hand, when the m-bit-th amplitude control signal Nvar[m] is 0, the control circuit 10 controls the control signal p[m] such that the switch SW_m is connected to the node of 0 V irrespective of whether the digital random signal Vdran is +1 or −1.

When k bits of m-bit amplitude control signals Nvar are 1, k switches out of the switches SW_1 to SW_m are connected to the positive reference voltage +Vref or the negative reference voltage −Vref based on Vdran=+1 or −1. As a result, the output Vo of the DA converter dndac (or the threshold input of the second comparator) is a voltage described below that is obtained by adding a fluctuation range ±Ve to the threshold voltage bn. Vo=bn±Ve Ve={kC/(mC+Cc)}Vref

Herein, k is the number of ones of Nvar, and m is the number of the capacitances 1C.

When the input terminal capacitance Cc is ignored because it is very small, the output Vo is as follows: Ve=(k/m)*Vref Vo=bn±Ve=bn±(k/m)*Vref

That is, the output Vo is the voltage obtained by adding the fluctuation value±Ve having the same sign as that of the digital random signal Vdran to the threshold bn, and the fluctuation range (amplitude) is a voltage range obtained by multiplying the reference voltage Vref by the number of ones (k) of the amplitude control signal Nvar (k times). Consequently, the first DA converter dndac converts the digital random signal Vdran generated based on the thermal noise to the analog random signal Ve. The amplitude of the analog random signal Ve can be varied with the amplitude control signal Nvar.

As illustrated in FIG. 12, in synchronization with the second clock CK2, the second comparator dn2 compares the analog output z_(n) obtained by converting the digital output z_(nd) of the digital adder an with the input voltage Vo=bn±Ve on which the noise having a predetermined amplitude is superimposed, and sets the output q_(n) to +1 when z_(n)>Vo, and sets the output q_(n) to −1 when z_(n)<Vo. On the other hand, the first DA converter dndac generates Vo=bn when the sample clock CK_(sam)=1, and generates Vo=bn±Ve when the sample clock CK_(sam)=0.

Consequently, as illustrated in FIG. 12, when the second clock CK2 rises from 0 to 1 during the hold mode when the sample clock CK_(sam)=0, the second comparator dn2 compares the analog input z_(n) with the threshold input signal Vo=bn±Ve. The threshold input signal Vo is an analog random signal obtained by adding the ransom fluctuation component Ve, which has the sign of the digital random signal Vdran and the amplitude by the amplitude control signal Nvar, to the threshold voltage bn serving as the reference. The second comparator dn2 sets the output q_(n) to +1 when z_(n)>Vo=bn±Ve, and sets the output q_(n) to −1 when z_(n)<Vo=bn±Ve.

As illustrated in FIG. 12, in the case where the analog input has the voltage of z_(n_2) (in case of |Ve|>|Zn|), the output q_(n) is q_(n)=−1 when the analog random signal Vo=bn+Ve, and the output q_(n) is q_(n)=+1 when the analog random signal Vo=bn−Ve. That is, when |Ve|>|Zn|, the output q_(n) of the second comparator dn2 is −1 or +1 depending on whether the sign of the fluctuation component Ve is + or − based on the digital random signal Vdran, and its probability is 50%.

On the other hand, in the case where the analog input z_(n) has the voltage of z_(n_1) (in case of |Ve|<|Zn|), the output q_(n) is q_(n)=+1 irrespective of whether the sign of the fluctuation component Ve is + or −. Conversely, in the case where the analog input z_(n) has the voltage of z_(n_3) (in case of |Ve|<|Zn|), the output q_(n) is q_(n)=−1 irrespective of whether the sign of the fluctuation component Ve is + or −. That is, when |Ve|<|Zn|, the output q_(n) of the second comparator dn2 is +1 or −1 without corresponding to whether the sign of the fluctuation component Ve is + or −, and its probability is 100%. This means the following description.

FIGS. 13A, 13B, and 13C are views illustrating the output function of the second comparator. In each of three graphs in FIGS. 13A, 13B, and 13C, the horizontal axis corresponds to the analog input z_(n), and the vertical axis corresponds to the probability that the output q_(n) of the comparator is q_(n)=+1. FIG. 13A illustrates the probability that the output is q_(n)=+1 in the case where a typical comparator compares the input voltage z_(n) with the threshold voltage bn (=0 V). In this case, the probability that the output is q_(n)=+1 is 1.0 (100%) when z_(n)>bn, the probability that the output is q_(n)=+1 is 0.0 (0%) when z_(n)<bn, and the probability that the output satisfies q_(n)=+1 is 0.5 (50%) when z_(n)=bn.

On the other hand, FIG. 13B illustrates the probability that the output of the comparator is q_(n)=+1 in the case where the above-described second comparator dn2 compares the input z_(n) with the analog random signal Vo=bn±Ve. In this case, the probability that the output is q_(n)=+1 is 1.0 (100%) when z_(n)>bn+Ve, and the probability that the output is q_(n)=+1 is 0.0 (0%) when z_(n)<bn−Ve. That is, in the case where |Zn|>|Ve|, the probability that the output is q_(n)=+1 is 1.0 (100%) or 0.0 (0%). However, when |Zn|<|Ve|, the probability that the output is q_(n)=+1 is 0.5 (50%). That is, the output is q_(n)=+1 or −1 with the probability of 50%.

FIG. 13C illustrates the output function (broken lines) in the case where the value of the amplitude control signal Nvar of the second comparator dn2 is reduced. When Navar is reduced, the fluctuation component Ve is reduced. As a result, a range in which the probability that the output is q_(n)=+1 is 0.5 (50%) is narrowed from −Ve1 to +Ve1 indicated by a solid line to −Ve2 to +Ve2 indicated by the broken line. That is, it is possible to arbitrarily change the range in which the probability that the output is q_(n)=+1 is 0.5 (50%) using the amplitude control signal Nvar.

Among the three types of the sigmoid functions illustrated in FIG. 4, FIG. 13A corresponds to the sigmoid function having T=0.5 in FIG. 4, FIG. 13B corresponds to the sigmoid function having T=2 in FIG. 4, and the broken line in FIG. 13C corresponds to the sigmoid function having T=1 in FIG. 4. Consequently, this means that the output function of the second comparator dn2 can be set to the sigmoid functions having different characteristics illustrated in FIG. 4 by varying the amplitude control signal Nvar.

Second Embodiment

FIG. 14 is a view illustrating the configuration of the comparison circuit (comparison unit) in a second embodiment. In addition, FIG. 15 is a view illustrating the signal waveform of the comparison circuit (comparison unit) in the second embodiment. The comparison circuit (comparison unit) do in the second embodiment has n sets of digital random signal generators DRAN each of which has thermal noise generation circuit (Rnoi and Cnoi) and first comparison circuit dn1, and the first DA converters dndac that convert the digital random signals Vdran to the analog random signals Ve and vary the amplitudes of Ve respectively. To each of the first DA converters dndac_1 to dndac_n, the amplitude control signal Nvar is input in common. The n sets of analog random signals (voltages) Ve as the outputs of the n first DA converters dndac_1 to dndac_n are added to the threshold signal bn.

The n sets of digital random signal generators DRAN_1 to DRAN_n generate their respective digital random signals Vdran independently of each other. As a result, the number of combinations of the n sets of digital random signals Vdran[n] can be 2^(n), and the number of voltage levels of the total of the n sets of analog random signals Ve can be n+1, including the case where there is no Vdran indicative of +1 and there are n Vdran indicative of −1 to the case where there is no Vdran indicative of −1 and there are n Vdran indicative of +1. That is, the total voltage of the n sets of analog random signals Ve has the n+1 voltage levels, and it is possible to vary the n+1 voltage levels using the amplitude control signal Nvar. The n+1 voltage levels by the n analog random signals Ve are added to the voltage of the threshold signal bn.

The signal waveforms in FIG. 15 are examples of seven digital random signal generators and first DA converters, and the analog random signals |Ve_1| to |Ve_4| corresponding to eight voltage levels (four negative voltages and four positive voltages) are illustrated. The eight voltage levels correspond to the following eight combinations of seven digital random signals Vdran:

(−1, −1, −1, −1, −1, −1, −1)=−Ve_4

(+1, −1, −1, −1, −1, −1, −1)=−Ve_3

(+1, +1, −1, −1, −1, −1, −1)=−Ve_2

(+1, +1, +1, −1, −1, −1, −1)=−Ve_1

(+1, +1, +1, +1, −1, −1, −1)=+Ve_1

(+1, +1, +1, +1, +1, −1, −1)=+Ve_2

(+1, +1, +1, +1, +1, +1, −1)=+Ve_3

(+1, +1, +1, +1, +1, +1, +1)=+Ve_4

FIGS. 16A, 16B, and 16C are views illustrating the output function of the second comparator in the case where the number of sets of the digital random signal generators and the first DA converters is one, two, or three.

In FIG. 16A, in the case where the number of sets thereof is one, there are two kinds of the digital random signals Vdran −1 and +1, and there are two kinds of the analog random signals −Ve1 and +Ve1. Consequently, the probability that the output q_(n) is q_(n)=+1 is 0.0 when the input z_(n) is z_(n)<−Ve, the probability is 0.5 when the input z_(n) is −Ve<z_(n)<+Ve, and the probability is 1.0 when the input z_(n) is +Ve<z_(n).

In FIG. 16B, in the case where the number of sets thereof is two, the number of combinations of the two digital random signals Vdran is 2²=4, which includes (−1, −1), (+1, −1), (−1, +1), and (+1, +1), and there are three kinds of the analog random signals −2Ve2, 0, and +2Ve2. Consequently, the probability that the output q_(n) is q_(n)=+1 is 0.0 when the input z_(n) is z_(n)<−2Ve, the probability is 0.25 (=¼) when the input z_(n) is −2Ve<z_(n)<0, the probability is 0.5 when the input z_(n) is z=0, the probability is 0.75 (=¾) when the input z_(n) is 0<z_(n)<+2Ve, and the probability is 1.0 when the input z_(n) is +2Ve<z_(n) is 1.0.

In FIG. 16C, in the case where the number of sets thereof is three, the number of combinations of the three digital random signals Vdran is 2³=8, and there are four kinds of the analog random signals −3Ve3, −Ve3, +Ve3, and +3Ve3. Consequently, the probability that the output q_(n) is q_(n)=+1 is 0.0 when the input z_(n) is z_(n)<−3Ve, the probability is 0.125 (=⅛) when the input z_(n) is −3Ve<z_(n)<−Ve, the probability is 0.5 (= 4/8) when the input z_(n) is −Ve<z_(n)<+Ve, the probability is 0.875 (=⅞) when the input z_(n) is +Ve<z_(n)<+3Ve, and the probability is 1.0 when the input z_(n) is +3Ve<z_(n).

As described above, according to the second embodiment, by increasing the number of sets of the digital random signal generators and the first DA converters, it is possible to make the waveform of the output function of the second comparator gradual to form a more natural waveform close to that of the sigmoid function.

[Annealing Operation of Boltzmann Machine]

In the Boltzmann machine described above, the digital random signal Vdran that randomly changes the sign of the noise based on the thermal noise is generated, the first DA converter dndac converts the digital random signal to the analog random signal +Ve or −Ve and varies its amplitude using the amplitude control signal Nvar, and the second comparator dn2 compares the input z_(n) with the threshold Vo=bn±Ve obtained by adding the analog random signal ±Ve to the threshold signal bn and generates the output q_(n) (+1 or −1). In addition, as in the second embodiment, in the case where a plurality of the sets of the digital random signal generators and the first DA converters are provided, the number of kinds of the analog random signal is increased, and the waveform of the output function of the second comparator dn2 is made similar to the more natural waveform close to that of the sigmoid function.

In the present embodiment, in the process of repetition of the execution of the Boltzmann machine, the largest value of the amplitude control signal Nvar is changed to the smallest value, and the largest value of the amplitude of the noise level ±Ve of the threshold of the second comparator dn2 is changed to the smallest value. With this, the level of the thermal noise is successively reduced, and the probability that the energy of the network achieves the optimal solution as the smallest value is increased.

FIG. 17 is a flowchart illustrating the operation of the Boltzmann machine in the present embodiment. In the Boltzmann machine in FIG. 17, first, the initial value of the weight w_(xy) is set to the fixed value of the model (S1), and the spin q_(i) of each node is set to an appropriate value (+1 or −1) (S2). Subsequently, the amplitude control signal Nvar of the analog random signal ±Ve is set to the largest value (S3). In this state, the Boltzmann machine is executed any number of times (S4).

FIG. 18 is a flowchart illustrating the detail of the execution step S4 of the Boltzmann machine. In the execution of the Boltzmann machine, the node or nodes serving as the operation target is or are randomly determined (S41), the comparator (comparison unit) d_(i) of the determined node or nodes is or are caused to operate, and the new output q′_(i) is or are generated from the analog outputs z_(i) of the digital adders a_(i) for the outputs q₁ to q_(n) of all of the current nodes (S42). As a result, the combination of the outputs q₁ to q_(n) of all of the nodes is changed to a different combination thereof by updating, and the digital adders a1 to an of the individual nodes generate addition values z_(1d) to z_(nd) corresponding to the new outputs q₁ to q_(n). Further, the energy value of the network to the new combination of the outputs q₁ to q_(n) transitions to another value.

In this transition, the second comparator operates in the following manner:

the second comparator outputs q′_(i)=+1 when Σ_(j)w_(ij)q_(j)−(b_(i)±Ve)>0, and outputs q′_(i)=−1 when Σ_(j)w_(ij)q_(j)−(b_(i)±Ve)<0.

The operation of the second comparator has the case where the sign of the output q_(i) is inverted and the case where the sign thereof is not inverted. In the case where the sign thereof is inverted, the energy value E of the network is reduced. Further, there are cases where, by adding the noise to the threshold, the sign of the output q_(i) is not inverted even in the operation that usually inverts the sign thereof and, in those cases, the energy value E of the network is increased. With the malfunction of the second comparator resulting from the noise, it is possible to transition to the combination of the outputs q_(i) generating the energy value that allows the escape from the minimum value as the local solution. Note that, during the execution of the Boltzmann described above, the calculation of the energy value of the network is not performed usually.

When the number of times of execution of the Boltzmann machine is assumed to be M times, and the number of times of reduction of the noise level of the annealing is assumed to be 3 times, for example, the random determination of the node of nodes serving as the operation target (S41) and the generation of the new output q′_(i) by causing the comparator d_(i) of the determined node or nodes to operate (S42) are repeated until the number of times of the execution thereof reaches M/3 times. Herein, causing the comparator d_(i) serving as the comparison unit of the node to operate means causing the second comparator d_(i) 2 in the comparison unit d_(i) to execute the comparison operation in synchronization with the clock CK2. The first comparator d_(i) 1 in the comparison unit d_(i) performs the comparison operation with the clock CK1 that is independent of the operation timing in the step S42, and continuously generates the digital random signal Vdran.

Returning to FIG. 17, the value of the amplitude control signal Nvar is reduced by any value (S5), and the above steps S4 and S5 are repeated while the total number of times of the execution does not reach M times (NO in S6). That is, the value of the amplitude control signal Nvar is reduced by any value, the magnitude of the noise amplitude Ve of the threshold voltage Vo=bi±Ve of the second comparator d_(i) 2 is reduced, and the step S4 is executed M/3 times again.

When the total number of times of the execution reaches M times (YES in S6), that is, when the amplitude control signal Nvar reaches the smallest value serving as an end value, the execution of the Boltzmann machine is ended, and data analysis processing is performed (S7). In the data analysis processing, the energy E of the network of the Boltzmann machine is calculated and recorded in a memory appropriately.

When the energy E has not reached the optimal value as the smallest value (or is not reduced to a value close to the optimal value), the number of times of the execution is increased or the reduction width of the amplitude control signal Nvar is reduced, and the flowchart in FIG. 17 is executed again. When the energy E has reached the optimal value (or is reduced to a value close to the optimal value), the execution of the Boltzmann machine is ended. For reference, when the number of times of the execution of the Boltzmann machine is infinite, the energy E reaches the smallest value (optimal value) theoretically.

FIGS. 19A, 19B, and 19C are views illustrating three phases of the execution of the Boltzmann machine in the present embodiment. Graphs on the left are the same as those in FIG. 3. On the other hand, each of graphs on the right denotes the sigmoid function of the second comparator d_(i) 2 of the present embodiment and the range of the output z_(i) of the digital adder a_(i) in the node ND_i. The sigmoid function corresponds to the case of three sets of the digital random signal generators and the first DA converters in FIG. 16C. Consequently, the sigmoid function provides different probabilities with the voltages of the analog random signal of −3V3 e, −Ve3, +Ve3, and +3Ve3.

In the state of FIG. 19A, the amplitude of the noise has the largest value, the characteristic of the sigmoid function is very gradual, and the probability that the output q_(i) of the second comparator is +1 is 0.5 only in the range of the output z_(i) of the digital adder a_(i). As a result, the second comparator generates the wrong output q_(i) with the probability of 0.5, and the energy of the network can be prevented from remaining at the local minimum value and move toward the optimal value.

In the state of FIG. 19B, the amplitude of the noise is a medium amplitude, the characteristic of the sigmoid function is slightly steep, and the probability that the output q_(i) of the second comparator is +1 is 0.125, 0.5, or 0.825 in the range of the output z_(i) of the digital adder a_(i). As a result, there are cases where the second comparator generates the wrong output q_(i) with the probability of 0.125, 0.5, or 0.825, and the energy of the network can be prevented from remaining at the local minimum value.

In the state of FIG. 19C, the amplitude of the noise is smallest, the characteristic of the sigmoid function is very steep, and the probability that the output q_(i) of the second comparator is +1 is 0.0, 0.125, 0.5, 0.825, or 1.0 in the range of the output z_(i) of the digital adder a_(i). As a result, the probability that the second comparator generates the wrong output q_(i) is reduced, and the energy of the network is likely to remain at the minimum value.

Thus, by varying the amplitude control signal Nvar from the largest value to the smallest value, it is possible to execute the Boltzmann machine while performing the annealing.

[Information Processing Device Having Boltzmann Machine]

FIG. 20 is a view illustrating an information processing device that has the Boltzmann machine. The information processing device is a computer, and has a processor CPU, a memory RAM, an arithmetic circuit unit 20 serving as the Boltzmann machine, an interface IF, an auxiliary storage device 21 such as a hard disk, an input/output device 22 such as a display and a keyboard, a network interface 23, and a bus 24 that connect the above elements.

The arithmetic circuit unit 20 is a network circuit that has circuit units of a plurality of the nodes in FIG. 5 and a plurality of digital arithmetic units between the nodes. When necessity for the processor CPU to determine the optimal solution occurs during execution of a program in the auxiliary storage device 21, the processor executes the arithmetic circuit unit 20 to acquire the optimal solution instead of executing the program to determine the optimal solution.

Thus, the Boltzmann machine in the present embodiment generates the digital random signal from the thermal noise with the small circuit scale, determine the sign of the noise in accordance with the digital random signal, varies the amplitude of the noise using the amplitude control signal Nvar, and generates the noise signal. Consequently, the Boltzmann machine generates the noise signal with the small circuit scale and easily varies the noise amplitude, and hence it is possible to perform the annealing control of the Boltzmann machine.

All examples and conditional language provided herein are intended for the pedagogical purposes of aiding the reader in understanding the invention and the concepts contributed by the inventor to further the art, and are not to be construed as limitations to such specifically recited examples and conditions, nor does the organization of such examples in the specification relate to a showing of the superiority and inferiority of the invention. Although one or more embodiments of the present invention have been described in detail, it should be understood that the various changes, substitutions, and alterations could be made hereto without departing from the spirit and scope of the invention. 

What is claimed is:
 1. A Boltzmann machine, comprising: a plurality of circuit units each of which has an adder that performs addition of a plurality of weighted input signals and a comparison unit that compares an output signal of the adder with a threshold signal to output a binary output signal; and a plurality of digital arithmetic units each of which generates each of the weighted input signals by weighting the binary output signal of a first circuit unit of a pair of the circuit units in the plurality of circuit units with a weight to output the weighted input signal to a second circuit unit of the pair of the circuit units, wherein the comparison unit (d_(n)) has: a first comparator (d_(n1)) that compares a thermal noise with a reference voltage in synchronization with a first clock to output a binary digital random signal; a first digital/analog converter (d_(ndac)) that converts the binary digital random signal to an analog random signal (V_(e)) that has two levels corresponding to the binary digital random signal, and varies a magnitude of the two levels of the analog random signal; and a second comparator (d_(n2)) that compares the analog random signal (V_(e)) combined with a fixed signal (b_(n)), as the threshold signal (V_(o)), to the output signal (Z_(n)) of the adder and outputs the binary output signal with a predetermined probability function.
 2. The Boltzmann machine according to claim 1, wherein a part of the plurality of circuit units that is randomly selected operates a predetermined number of times while an amplitude of the analog random signal is successively reduced.
 3. The Boltzmann machine according to claim 1, wherein the first digital/analog converter controls a sign of the analog random signal based on the binary digital random signal, and varies an amplitude of the analog random signal based on a magnitude control signal that is supplied to the first digital/analog converter.
 4. The Boltzmann machine according to claim 3, wherein the first digital/analog converter has a plurality of capacitive elements and a plurality of switches provided to the plurality of capacitive elements respectively, each of the plurality of switches applies a positive reference voltage or a negative reference voltage based on the binary digital random signal to a first terminal of each of the plurality of capacitive elements, and generates the analog random signal at a second terminal common to the plurality of capacitive elements, a number of the plurality of switches applying the positive voltage or the negative voltage is controlled by the magnitude control signal.
 5. The Boltzmann machine according to claim 1, wherein the second comparator performs a comparison operation in synchronization with a second clock, and outputs the binary output signal.
 6. The Boltzmann machine according to claim 3, wherein the comparison unit has a plurality of sets of first comparators and first digital/analog converters that receive binary digital random signals of the first comparators respectively, the magnitude control signal is input in common to each of the first digital/analog converters, and analog random signals of the first digital/analog converters are added up to input to the second comparator.
 7. The Boltzmann machine according to claim 1, wherein the output signal of the adder is a digital output signal, each of the circuit units has a second digital/analog converter that converts the digital output signal of the adder to an analog output signal, and the comparison unit compares the analog output signal with the threshold signal.
 8. A method for controlling Boltzmann machine, the method comprising: adding, in a plurality of circuit units in the Boltzmann machine, each of which has an adder and a comparison unit, a plurality of weighted input signals; comparing, in the comparison unit in each of the circuit units, an output signal of the adder with a threshold signal to output a binary output signal; generating, in a plurality of digital arithmetic units in the Boltzmann machine, the weighted input signals by weighting the binary output signal of a first circuit unit of a pair of the circuit units in the plurality of circuit units with a weight to output the weighted input signal to a second circuit unit of the pair of the circuit units; comparing in a first comparator of the comparison unit in each of the circuit units, a thermal noise with a reference voltage in synchronization with a first clock to output a binary digital random signal; converting in a first digital/analog converter of the comparison unit in each of the circuit units, the binary digital random signal to an analog random signal that has two levels corresponding to the binary digital random signal, with variation in a magnitude of the two levels of the analog random signal; and comparing in a second comparator of the comparison unit in each of the circuit units, the analog random signal combined with a fixed threshold signal to the output signal of the adder to generate the binary output signal with a predetermined probability function.
 9. An information processing device comprising: a processor that executes instructions of a program; and an arithmetic circuit unit that operates in response to an operation request from the processor, wherein the arithmetic circuit unit includes a plurality of circuit units each of which has an adder that performs addition of a plurality of weighted input signals and a comparison unit that compares an output signal of the adder with a threshold signal to output a binary output signal; and a plurality of digital arithmetic units each of which generates each of the weighted input signals by weighting the binary output signal of a first circuit unit of a pair of the circuit units in the plurality of circuit units with a weight to output the weighted input signal to a second circuit unit of the pair of the circuit units, wherein the comparison unit has: a first comparator that compares a thermal noise with a reference voltage in synchronization with a first clock to output a binary digital random signal; a first digital/analog converter that converts the binary digital random signal to an analog random signal that has two levels corresponding to the binary digital random signal, and varies a magnitude of the two levels of the analog random signal; and a second comparator that compares the analog random signal combined with a fixed threshold signal to the output signal of the adder to generate the binary output signal with a predetermined probability function.
 10. The information processing device according to claim 9, wherein a part of the plurality of circuit units that is randomly selected operates a predetermined number of times while an amplitude of the analog random signal is successively reduced. 